What is a switching amplifier
Switching and controlling with transistors I
Table of Contents
- 1 Introduction
- 1.1 Radio timer module SC-77-M from Conrad
1.2 Switching with transistors has priority
- 2.1 Exotic circuit for low voltage drop
4. Switching with a complementary Darlington pair
5. Switching module and switching amplifier
6. Switching amplifier, the first solution
7. Alternative: MOSFET switching amplifier
8. First solution with MOSFET
- 8.1 Long line problem
10. A better SC77M circuit
11. Link list
The trigger for this electronics mini-course was an email from an electrical engineering student who asked for advice on how to amplify the switching output of a certain time switch module and how to connect it correctly so that it can be used to control a "strong" relay that is operated by a 12- VDC car battery turns a car headlight on and off in order to illuminate any object with spotlight. Battery operation and low voltage are suitable because where this is to be used, there is no 230 VAC mains connection. The student had problems with it because the enclosed application notes for the module did not lead to the desired success. What was worrying for me was that the student with the simple circuit problem was already in his sixth semester, so he had problems realizing such a simple circuit himself without outside help. However, my concerns did not and do not go to the address of the student ...
If you want to comment on this topic because someone has had a similar experience or is aware of a similar case, please write me an email with the subject "Too much theory and (almost) no practice". E-mail address on the index page at the bottom left. I look forward to every response!
1.1 Radio timer module SC-77-M from Conrad
The radio time switch module SC-77-M from Conrad Electronics was used. It must be said that this project was topical in 2004. In October 2006 this radio timer module was no longer available! However, the content about switching with transistors is a universal topic. With the knowledge of these basics and examples, each reader can make appropriate adjustments to other modules. Temperature switching modules or modules for completely different physical measurements can also be used.
The technical data enclosed with the radio time switch module SC-77-M was (are) incomplete. With regard to the switching output, one only reads that it is a transistor stage with an open collector and that active low applies. So you know that the time switch is either switched on or off (???) when the NPN transistor integrated in the time switch is switched on. The emitter is connected to GND. But nowhere do you read how much collector current this transistor can withstand and how large the maximum open collector-emitter voltage can be. This is information that should actually be taken for granted.
At the time, I assumed that customer complaints would result in a follow-up check by Conrad. Indeed, Conrad responded with an extra sheet with an additional application note with the excuse that an error had crept in on page 7 of the original data sheet. The corrected circuit still has defects: The base current of the following external NPN transistor is only 0.25 mA, which in switching mode (saturated state of the transistor) is only sufficient for a collector current of less than 10 mA and that is usually too little for a relay whose coil is operated at 12 VDC. The current is more in the order of magnitude between 30 and 100 mA. In the switched-on or saturated state, a gain of a maximum of 20 to 30 with a collector current of about 100 mA should be assumed for low-power transistors, otherwise the collector-emitter voltage will be too high. Then the NPN open collector output stage (switching output) also controls an NPN transtor switching stage. This means that the relay is switched on when the open collector transistor inside the watch module is open, i.e. in the inactive state. So much for the technical information about the radio time switch module SC-77-M from Conrad ...
ATTENTION: Important section !!!
Now, however, it is the case that Conrad has a somewhat strange nature active LOW To be defined: It applies here that the switching output has a LOW level when the timer function is inactive. The relay must therefore be picked up when the internal switching transistor is switched off (passive), i.e. does not supply the LOW level. This is confusing, although there is also a definition that just refers to the logic level. In the book Semiconductor circuit technology by U.Tietze and Ch. Schenk can be read in the chapter Basic logic circuits: "Since the outputs are only low-ohmic in the LOW state, they are also referred to as active low outputs." Since I am assuming that there are other switching modules and time switch modules in which the meaning of the active designation corresponds to the definition of the circuit logic and the active state of the output, e.g. when a lamp is lit or a pump motor is running, it is only at the end This electronics mini-course specifically deals with the above-mentioned time switch module SC77M from Conrad. Almost all switching amplifiers shown here can just with any other switching modules with open collector NPN transistor or open drain MOSFET output stages can be used, which one real Have an active low output and this does not include the Conrad time switch module SC77M! In Chapter 10 "A better SC77M circuit"Two extra circuits follow in Fig. 14 and Fig. 15 with a description for the mentioned Conrad product.
1.2 Switching with transistors has priority
This electronics mini-course begins with learning how to switch with bipolar transistors, with Darlington switching stages and also with low-power MOSFETs, before finished circuits are shown and explained. The small study of this electronics mini-course serves on the one hand to learn what is important when switching with these transistors and on the other hand with such knowledge one is simply not dependent on insufficient application instructions in the enclosures of any time switches or other switching modules. The more independence the electronics trainee develops, the better it is for him and that is always the credo of my electronics mini-courses in the electronics compendium.
We will first deal with transistor switches which are constructed with bipolar transistors, whereby the Darlington is also discussed and the use of low-power MOSFETs is also discussed and we consider the advantages and disadvantages between bipolar transistors and MOSFETs . It is obvious that this presupposes that one knows how these transistors basically work. There is also some teaching material from Patrick Schnabel in the ELKO. Please refer to the list of links on the subject of transistors (bipolar and MOSFETs).
In the following, we limit ourselves to low-power transistors in the collector (bipolar) or drain current range (FET) of 100 mA or a little more and with a maximum power dissipation of around 500 mW. BC547C, BC550C (NPN) and BC557C and BC560C (PNP) are recommended as very inexpensive bipolar transistors. The letter at the end of the number indicates the current gain range. It hardly makes a difference in price, but makes a significant difference in quality to choose the C-type, if one is available. So e.g. BC547C instead of BC547B instead of BC547A. You can find out more about this on the bipolar transistor page (see table on NPN / PNP transistors).
A few types are discussed as low-power MOSFETs. There is the well-known N-channel MOSFET BS170 with a maximum drain current of 270 mA, a drain-source resistance of 5 ohms and a maximum power dissipation of 625 mW. In terms of current and power, this MOSFET is also called the MOSFET counterpart to the bipolar NPN transistors mentioned above. But there is also a P-channel MOSFET that is a counterpart to the PNP transistors mentioned. It is the BS250 with a maximum drain current of -230 mA, a drain-source resistance of 14 Ohm and a maximum power dissipation of 700 mW. You can already see something fundamental here: In the production of MOSFETs, it is easier to get low-ohm drain-source resistances with N-channel types than with P-channel types. A few other low-power MOSFETs with significantly lower drain-source resistances (RDS_on) are also discussed in the corresponding chapter. These are BSS295 (0.3 Ohm) and 2SK2961 (0.26 Ohm) as N-channel types and VP0300LS (2.5 Ohm) and VP0800L (5 Ohm) as P-channel types. IRLD024 (N-channel, 0.1 Ohm, 2.5 A, 60 V) and IRFD9024 (P-channel, 0.28 Ohm, 1.6 A, 60 V) in the DIL housing are also interesting and worth mentioning. When the drain-source resistance is mentioned, the switched on state is always meant!
2. Switching with NPN and PNP transistors
Figure 1 shows the simplest method of electronic switch with NPN and PNP transistors. In both fields, R2 is initially missing, which is connected in parallel to the base and the emitter. Part 1.1 shows an NPN switching stage. If Ue is at LOW level, i.e. the voltage is clearly below the base-emitter threshold voltage of T, T is open and Ua, when unloaded (without RL), the voltage of + Ub. This is the HIGH level. The source resistance corresponds to the value of R3, which is easy to see, because if the load resistance RL is the same as R3, both resistors act as voltage dividers with equivalent resistances, which halves the voltage at Ua (Ub / 2). If Ue has a HIGH level, i.e. a voltage greater than the base-emitter threshold voltage of T, R1 must be dimensioned in such a way that the base current Ib controls the transistor into saturation. Although the current amplification factor of a small-signal transistor can be specified as far more than 200 in the data sheet, one should not expect more than 20 to 30 if T is to work in saturated operation and the collector current is in the lower 100 mA range. With high-current transistors (e.g. BD239 or 2N3055-Oldy) and in the ampere range, you have to go down to a factor of 10 in saturated operation. If you choose a factor that is too high, saturation is no longer guaranteed and the collector-emitter voltage increases in an impermissible manner. In the case of the NPN and PNP transistors specified above, the collector-emitter saturation voltage with a collector current of 100 mA is approximately 0.2 V. This is the LOW level voltage at Ua. The source resistance is extremely low at LOW level because of the switched on transistor. In order to better understand the relationship between collector-emitter voltage, collector and base current, consider the output characteristic field on the ELKO page transistor characteristic fields by Patrick Schnabel.
If you add R2, R1 and R2 form a voltage divider. This serves to increase the switching threshold above that of the transistor in the + Ub direction. This can be used to improve the signal-to-noise ratio. It is important that the cross current through R1 and R2 is three times as high as the base current should be. The greater this ratio, the sharper the kink from the blocking to the conducting or saturated transistor.
Part 1.2 is basically the same as part 1.1, but with some opposite signs. Instead of an NPN transistor, a PNP transistor is used. A current flows through the base of the transistor when Ue is at LOW level, whereby a collector current also flows. The reverse is also true here that the output resistance is then etrem low when the HIGH level is at Ua. RL is not a significant burden. If there is a LOW level at Ua, the output resistance corresponds to the value of R3. However, this only has a disadvantageous effect if Ua were referred to + Ub via RL. R1 and R2 have the same purpose as with the NPN switching stage in part 1.1: The switching threshold is increased above that of the transistor, which here, however, means that the switching threshold at Ue is shifted towards GND.
The switching threshold Ues is calculated for drawing file 1.1 ...
Ues = ((R1 / R2) + 1) * 0.65V
... and for drawing file 1.2
Ues = Ub - (((R1 / R2) + 1) * 0.65V)
The voltage of about 0.65 V is the base-emitter threshold voltage of the transistors in the two circuits.
2.1 Exotic circuit for low voltage drop
Of course we don't leave anything out, not even anything exotic. Something that hardly anyone knows: You can also use the transistors the wrong way round, i.e. swap collector and emitter, and that actually works. This circuit in the NPN (part 2.1) or in the PNP version (part 2.2) has the advantage that the collector-emitter saturation voltage is much lower. Instead of around 200 mV, it has a value in the 10 mV range. Great transistor switch, isn't it? The reader is already wondering how stupid I actually am to organize such a large "theater" around picture 1 when the circuit diagrams in picture 2 show the ideal and ultimate transistor switches. Well, they have their very own disadvantages, almost something like a personal touch and these are very annoying: Disadvantage 1 is that + Ub must always be lower than the maximum permissible emitter-base voltage of the transistors (vice versa!) . Unfortunately, this voltage value is always very low at 5 to 7 V. Disadvantage 2 is that the transistor in this circuit simply refuses to amplify its current. The value is less than 1. You can only say: nothing except expenses! :-)
3. Switching with NPN Darlington pair
In parts 3.1 to 3.22 we consider what happens when switching using NPN Darlington circuits. The same applies if PNP transistors are used, but with the opposite sign in voltage and current. That is why the use with PNP transistors is not dealt with additionally. If you don't know what Darlingtons is about, first look in the link list and read the corresponding links.
Part 3.1 illustrates a linear Darlington circuit in a collector circuit with a relatively high current gain and a voltage gain of just under 1, although this only applies to relatively small voltage changes at Ue. This circuit is also called emitter follower. If we consider the DC voltages, Ua is always lower than Ue by the base-emitter threshold voltage of T1 and the same of T2 if both transistors are working correctly and this is the case when Ue never exceeds the value of + Ub. If Ue remains a few volts below + Ub, then both transistors develop their maximum current gain values because the collector-emitter voltages are then sufficiently high. Take a look at the relevant data sheets for transistors. The base current Ib1 results automatically from the collector current Ic2 divided by the product of the current gains of T1 and T2.
We now switch to drawing file 3.2 with the same Darlington step. However, the emitter of T2 is connected to GND. It is an emitter-connected Darlington stage with the typical voltage and current amplifying effect. A current-limiting resistor is required in the base line of T1, labeled R1. We think of R2 away at first. Ue is initially at LOW level. This means that the voltage must be clearly lower than the sum of the two base-emitter threshold voltages of T1 and T2 of around 1.3 V. At 1 V, with a very high current gain (ßT1 * ßT2), a low collector current Ic2 can already flow. It is therefore advisable to set the LOW level voltage to the lowest possible value close to the GND level. If you want a higher switch-on level, R2 must be added. R1 and R2 then act as voltage dividers in the same way as described in chapter "Switching with NPN and PNP transistors"(Picture 1) is described.
The voltage difference between + Ub and Ua divided by R3 gives the collector current Ic2. If Ue is at HIGH level and this voltage is sufficiently above the sum of the two base-emitter threshold voltages of T1 and T2, Ue minus these two voltages divided by R1 determines the base current Ib1 - without the presence of R2. With R2, Ib1 is subtracted by the current through R2. Here, too, the more "sharp-edged" the transition between non-conductive and conductive, or open and closed Darlington switch, the greater the cross-current Iq should be selected in relation to Ib1. Ib1 must be so high that the collector-emitter voltage of T1 is as low as possible. This can be in the range of a few tens of millivolts if the collector current of T1 is only a few mA. This is exactly the case when R3 is the coil resistance of a relay that causes a current (Ic2) of around 100 mA.
Ua, however, can never be smaller than the base-emitter threshold voltage of T2, and this is illustrated in part T3.21. T1 is shown here symbolically as a potentiometer between the collector and the base of T2. If T1 becomes very saturated, it is comparable to setting the potentiometer at a very low resistance, e.g. so that only a few tens of millivolts are dropped across it. If you even short-circuit the potentiometer, this corresponds to the direct connection of collector and emitter of T1. This state is identical to a diode and the typical diode threshold voltage (part 3.22).From this we can understand that a Darlington stage can never have a lower output voltage at Ua than a diode threshold voltage. In our case with Ic2 in the range of 100 mA, this voltage is around 0.7 VDC, with high-power transistors (T2) and a few amps, including UceT1, it is around 1 VDC or a little more.
4. Switching with a complementary Darlington pair
Part 4.1 shows the linear operating mode of the complementary NPN Darlington pair. NPN, because the control function corresponds to that of an NPN transistor. Whether the complementary Darlington circuit as a whole has NPN or PNP properties is determined by the first transistor (T1), i.e. the one which is controlled from the outside at its base. The difference to part 3.1 is the minimum differential voltage Ue minus Ua. This is only one base-emitter threshold voltage, namely that of T1. This also applies to the circuit in section 4.2.
Ua / R determines the T2 collector current. The base current Ib1 results automatically from the collector current Ic2 divided by the product of the current gains of T1 and T2. Such complementary Darlingtons are ideal for voltage regulation circuits with a relatively low minimum dropout voltage (see link list).
Partial image 4.2 corresponds functionally to partial image 3.2. Still, there are differences. In part 4.2, the voltage of the switching threshold without R2 corresponds only to a base-emitter threshold voltage, namely that of T1. Is there a difference in terms of the minimum output voltage Ua (LOW level)? No, there is none if Ib1 is also dimensioned generously, i.e. is much larger than what is required with Ic2 divided by the product of the two current gains of T1 and T2. The same considerations apply to sections 4.21 and 4.22 as to sections 3.21 and 3.22: Here too, a Darlington stage can never have a lower output voltage at Ua than a diode threshold voltage.
We now come to a special consideration of Darlington connections in Figure 5.
The Darlington circuit in section 5.1 has been given the additional resistor R4. What's that good for? When Ue is LOW, T1 is open. Isolate the path base to emitter and collector to emitter. The result is that the emitter of T1 and thus the base of T2 "hang in the air". That is not entirely "clean", although you can talk yourself out of the fact that this resistance is also not included in integrated Darlingtons. But that is precisely a prejudice because there are Darlingtons built in with this resistance. This resistor, here R4, defines the base with the potential of the emitter of T2 when T1 is open. If T2 has a relatively high current gain in the linear operating range and the lead from T1 emitter to T2 base is relatively long, it can happen that capacitively coupled interference AC voltages could control this line T2. Unwanted collector current Ic2 occurs. R4 can be dimensioned so that about 1/10 of the T2 base current flows through it. The current through R4 is always constant because the voltage across it always has the value of the base-emitter threshold voltage of T2, provided that T1 is switched on.
R4 is also recommended for a completely different reason: If the Darlington stage is used for higher switching frequencies, it would switch on faster than it off without R4. The edge steepness of switching on would be significantly higher than when switching off. This is due to the fact that the T2 base is open when it is switched off and the charge carriers can only flow away with difficulty (parasitically). However, high-frequency switching requires that R4 must be relatively low-resistance and a capacitor must be connected in parallel to R1, which reduces the disturbing Miller effect, which results from the current amplification of T1 and T2 and the parasitic collector-base capacitance of T1, compensated. However, this topic would go beyond this electronics mini-course and would have to be dealt with separately. This is only about slow switching processes, such as switching a lamp or a relay. Relay control will be topic number 1 from the following chapter.
Part 5.2 shows where R4 is to be installed in the complementary Darlington stage. When T1 is open, the T2 base is also undefined. With R4 the base has emitter potential.
5. Switching module and switching amplifier
In the following figure 6 shows the switching output of a switching module in two circuit variants. It can also be a (radio) clock switching module, excluding, as already mentioned, the radio clock switching module SC77M from Conrad. Any number of modules such as alarm devices that react to physical limit values are also possible. Solid-state switching outputs with an open collector (bipolar NPN transistor) or an open drain (N-channel MOSFET) are often found. As already mentioned, it is important here that when the transistor is switched on, the external consumer must also be switched on, i.e. in the active state.
In the following sections, however, a special case is discussed. This case is perhaps more common than one might assume, namely when the data sheet enclosed with the purchased product provides incomplete information. The important information that is missing is described in detail here in the introduction.
The two following figures, Figure 6 and Figure 7, show possible solutions that should be used with caution. We'll see why ...
As can already be seen from the introduction, the aim is to switch a large current with a relay. A Siemens relay with an excitation coil of 12 VDC and a current of 83 mA was used for this. We already know that a bipolar low-power transistor in saturation mode and a current of around 100 mA can be expected to have a maximum current gain of 20 to 30.
Part 6.1 shows how the relay REL is controlled by the PNP transistor T2. Diode D, as a so-called freewheeling diode, serves the sole purpose of short-circuiting the high self-induction voltage when the relay coil is switched off in order to protect T2 from possible destruction. The selected current gain for switching operation, i.e. for a minimum voltage drop between emitter and collector of approx. 0.2 VDC, is 25. This requires a base current of 3.3 mA. With the selection of R3 = 2.7 k-Ohm, it is 3.6 mA. We now know that it is not "clean" if the base of T2 is open, which would be the case without R2, if T1 (open collector NPN transistor) of the switching module is open. With a value of 10 k-Ohm, the maximum current through R2 is 65 µA. It may also be less than stated above with 1/10 of the base current. It always plays a role to choose a reasonable resistance value and 10 to 100 k-ohms (or possibly even more) is quite reasonable, if you consider that the line to the T2 base is not unnecessarily long and not unnecessarily close to one There is a source of interference that could capacitively couple an interference voltage. There are also no high switching speeds which would require R2 to be low-resistance. Additional interference suppression measures follow in Chapters 6 and 8.
The base current is calculated as follows:
R1 = (Ubmin - 1V) / Ib
Ubmin of 10.8 VDC is a discharged 12V lead-acid battery. The voltage of 1 V is made up of the base-emitter threshold voltage of T2 and the collector-emitter saturation voltage of T1. Calculated, this gives a value of 2.7 k-Ohm for R1 from the 5% resistance series. The current ISA is about 3.6 mA. The current through R2 with only 65 µA is not taken into account in the calculation.
And now to the uncertainty, as described above: You may not know whether this transistor T1, which is integrated in the switching module, is already being subjected to too much current. If not, you can be satisfied with this solution, otherwise we come to the next step in part 6.2:
This circuit differs from part 6.1 only in the use of a PNP Darlington instead of a PNP transistor. This can consist of a single component, e.g. BC516, or two single transistors, e.g. 2 pieces BC560C. The current gain of a self-implemented BC560C Darlington stage ranges from 160,000 to 640,000 in linear operation. Nevertheless, one should not use more than about 1000 in saturated switching mode. This requires a base current Ib2 of 83 µA. As is well known, there is twice the base-emitter threshold voltage of about 1.3 V between the base and emitter of a Darlington. A current of 13 µA flows through R2. ISA stays below 0.1 mA, which T1 of any switching module will easily handle. If not, the switch module is simply worthless!
So everything would be fine with that, if appearances are not deceptive. We must not forget that if the 12V relay is operated on a car battery, the discharge voltage of which is 10.8 VDC (cell voltage = 1.8 VDC), the relay must still securely pick up. If there are no other significant voltage drops, most relays will just barely do so. But you have to get detailed information from the data sheet or try your hand at it. When using a darlington, + Ub drops from a minimum of 10.8 VDC to around 10 VDC. There could be significant tightening problems with certain relay types. This situation does not get any better if a complementary Darlington stage is used, as described above for switching operation (Figure 4) and Figure 7 illustrates:
6. Switching amplifier, the first solution
Figure 8 shows a good solution with two external transistors. The relay REL only suffers a voltage drop of about 0.2 VDC, which is caused by the collector-emitter saturation voltage of T3. Ib3 is again 3.6 mA as in part 6.1 Ib2. Because this current is perhaps too high for the NPN open collector transistor T1, there is an additional PNP transistor stage with T2 in the middle part. This inverts the control voltage and that is the reason why REL must be operated with an NPN transistor. Note the current arrows and it immediately becomes clear how T1 controls T2 and this controls T3. The current load of T1 at around 0.1 mA remains the same as in Figures 6 and 7.
The capacitor C and R5 are new here. This option is recommended if the circuit is used in an environment with strong electrical interference. C2 reduces the input impedance for AC voltages. The capacity of 100 nF is only a recommendation. When T1 switches off, the low-ohmic differential resistance of the base-emitter junction of T2 with R5 in series and the capacitance C forms a dominant low time constant, although R2 is connected in parallel to the resistor just mentioned. The lower Ib2 becomes due to the discharge of C, the higher the resistance of the base-emitter circuit and when Ib2 approaches zero, only the time constant C * R2 dominates. Further discharge is slower. The transition between the conductive and the blocking state of T2 is more gradual, the larger C is chosen. This also makes the switching process of the relay, if not to the same extent, creeping, which damages the contacts due to long-term sparking. Therefore, one should not overdo it with the capacity of C. With the specified value of 100 nF, the R2 * C time constant is 10 ms. Re R5: This resistor is only required if C is used. Without R5, the differential input resistance of T2 alone would be far too low.
7. Alternative: MOSFET switching amplifier
Instead of bipolar transistors, you can also switch with MOSFETs and this also has a fantastic advantage: MOSFETs are voltage-controlled (no input current!), Bipolar transistors are current-controlled. In order to understand this difference, among other things, it is necessary to familiarize yourself with the basics again, this time with MOSFETs. You can find out more about this in the link list.
Since we are only dealing with very low switching frequencies here, the often high gate-source capacitance of power MOSFETs is irrelevant. At high frequencies, this fact must be taken into account with appropriately powerful driver circuits (capacitive load), e.g. in switched-mode power supplies.
Figure 9 compares a bipolar NPN switching stage (part 9.1) with a switching stage with an N-channel MOSFET (part 9.2). In contrast to the current and voltage control, there is another essential difference. A bipolar transistor changes its collector-emitter saturation voltage insignificantly depending on the collector current, unless the dimensioned current gain is "maxed out", i.e. the parameters of the data sheet are adhered to. It's completely different with the MOSFET. If this works in switching mode, the resistance between drain (D) and source (S) is a "real" resistance. This means that the voltage drop across this resistor is linear to the drain or source current within certain permissible limits. Depending on the application, this difference can mean that the bipolar transistor or the MOSFET has the lower voltage drop in the main current path. The tables in Figure 9 give the necessary information based on an example:
The table in part 9.1 shows three bipolar transistors of roughly the same current and power class. With a correspondingly low selected current gain, all have a collector-emitter saturation voltage of about 0.2 V at a collector current Ic of 100 mA. You can exchange transistors of the same current / power class, the collector-emitter saturation voltage always remains about the same. This is very different when using MOSFETs, as the table in part 9.2 shows. With a current of 100 mA, the BS170 shows a voltage loss of 0.5 V. If this voltage drop is important, then one of the bipolar "colleagues" from Part 9.1 is clearly superior to this MOSFET. If we look at the MOSFET IRLD024, it only generates a voltage drop of 10 mV. The resistance and the voltage drop is thus 50 times lower. The IRLD024 is TTL controllable. This means that the TTL-HIGH level of typically 3.4 V at the gate of this FET already controls it fully.
What is the meaning of the resistor R1 at the entrance of the gate of the MOSFET? What use is this something, because it is a voltage control? On the one hand, the gate-source capacitance can destabilize a driver circuit, i.e. it can cause it to vibrate; on the other hand, the MOSFET circuit itself can vibrate at very high frequencies if it is directly controlled with very low resistance. In the case of high-frequency circuits, R1 must be chosen carefully, usually with a low resistance, so that R1 and the gate-source capacitance do not generate an interfering low-pass filter cutoff frequency that is too low. Instead of R1, small inductors are sometimes used. In our examples, which are always about slow switching processes, R1 can easily have a value of 1 k-ohm and you are therefore on the safe side with regard to stability. The same applies to R1 in sub-picture 10.2.
Figure 10 shows the same with a PNP transistor and a P-channel MOSFET. Here the two MOSFETs BS250 and VP0808L are inferior to the bipolar "colleagues" and the VP0300L is close. The IRFD9024 can stand up. If you compare the tables in Fig. 10 with Fig. 9, you get the impression that it is easier to obtain particularly low-resistance N-channel MOSFETs than P-channel MOSFETs. That is actually the case, which is noticeable when you look around in a semiconductor catalog. At INELTEC, an electronics trade fair that takes place in Basel every two years, I was once informed that it is more difficult to get low-resistance structures in P-channel MOS processes and that is why there are also so-called high-side MOSFET driver stages in which N- Channel MOSFETs are in use and so that the gate control voltage is high enough, it must be stepped up using a switched capacitor boost circuit.
8. First solution with MOSFET
Part 11.1 with the PNP switching stage is the repetition of part 6.1 and is the inspiration for the P-channel MOSFET method in part 11.2, whereby it is advisable to install the interference protection capacitor C due to the higher input resistance. With a value of 22 nF, this has a time constant of about 10 ms with R2 (compare with Figure 8). This is sufficient to effectively suppress medium-frequency interference caused by capacitive coupling. 50 Hz ripple voltages can also be effectively suppressed if the parasitic coupling capacitance is only very low. R3 serves the same purpose as R1 in partial figures 9.2 and 10.2 and has already been described there. R1 (part 11.1) usually does not need it and can be bridged. At best, it could be necessary if a MOSFET has an extremely low gate-source threshold voltage. However, if + Ub is higher than the maximum permissible gate-source voltage, which is usually ± 20 V, then R1 is required. In this case, R1 and R2 serve as voltage dividers to protect the gate from overvoltage. So it all depends on the application. You are largely free to choose R2. With 560 k-Ohm, the switching output of the switching module has a current of 22 µA.
8.1 Long line problem
Applications are conceivable in which a switching module is far away from the switching amplifier with the relay. So it takes a long line. Now one could come up with the idea of massively increasing C in order to effectively suppress interference from very long cables that act as antennas. As already mentioned, this can, however, delay the switching on and off of the relay in an impermissibly creeping manner.
The better solution is to use shielded cable, remembering that such a cable has a capacitance of around 200 pF / meter or more. Here C2 has a different purpose.Without C2 the relay would switch on very briefly when the operating voltage + Ub is switched on, because the cable capacitance would form a differentiating element with R2. First, the full gate-source voltage is applied and controls the MOSFET. The time constant Ck * R2 determines the switch-on duration of the relay, which is still very short with a cable of 20 meters, with a capacitance of more than 4 nF and R2 of 560 k-Ohm, at around 2 ms and the armature of the relay hardly picks up . But the matter is uncertain. With C = 22 nF this effect is significantly reduced, because in the brief instant of switching on + Ub Ck and C act as voltage dividers and in the present example the maximum gate-source pulse voltage is reduced to 1/5. This damping becomes even better if C is reduced to 100 nF and, in order to keep the R2 * C constant R2, to 100 k-Ohm, which then increases the current ISA to 0.1 mA and would still be beneficial for T1.
9. Second solution with MOSFET and NPN transistor
We remember the circuit in Figure 8 and replace the PNP transistor T2 with a P-channel MOSFET as shown in Figure 13 above. This has the advantage that you do not have to worry about current amplification for T2. The input resistance of the MOSFET is extremely high and is in the tera-ohm range, and this means, as in Figure 11.2, that the current ISA can in principle be dimensioned as low as desired. Here, too, it is defined with R2 = 560 k-Ohm to 22 µA.
With this mixture of MOSFET preamplifier and bipolar output stage, you are free to choose the components. There are enough bipolar PNP transistors with the properties mentioned above and used here and they are very cheap. There is also a choice for the MOSFET because the drain-source resistance is not critical in a wide range, as it only has to supply a current of a few mA. For example, the BS250 is suitable for T2 and the BC547B for T3. Both are very inexpensive.
10. A better SC77M circuit
As already mentioned, in the logic of the circuit, the active state of the transistor is not identical to the active state of the connected device (lamp, motor, etc.). However, it is not enough to use just one transistor to control the relay, because the gain of one transistor would have to be too high and that does not do justice to the saturated state of a switched transistor. T3 is extended by T2. Both together, however, intentionally do not form a Darlington, because this would lead to an impermissibly high voltage loss for the relay. Therefore the collector of T2 is connected to a resistor R1 with the voltage of + Ub. The base is also connected to R2 with high resistance to + Ub. The switching current at the switching output remains very low at around 0.1 mA. When T1 is switched on - active low in the logic of the circuit - the base of T2 is pulled to GND potential. This opens T2 and thus also T3. So that the base of T3 is not exposed, it is pulled to GND potential with R4. In this state the relay is de-energized. No consumer controlled by the relay contact is active.
The green LED serves as a voltage stabilizer with a voltage of around 2 VDC. The time switch module feeds this voltage via diode D3 and also keeps a nickel-cadmium or nickel-metal hydride battery in a floating charge. This battery supplies the operating voltage for the clock if + Ub fails. D3 blocks the flow of current back into the circuit. The operating voltage of the module is just over 1.2 VDC. This battery idea comes from the documents for the time switch module SC77M. This battery charging process has an interesting effect: if the battery is discharged, all of the current flows into this battery. In this discharged state, the voltage for the green LED is too low. It does not glow at all or only weakly. The brighter it gets over time, the more it signals that the battery is charged.
Figure 15 manages with the use of a low-power MOSFET with only one transistor. How this MOSFET circuit works should be derived from the entire course, be clear and need no additional explanation. The drain-source voltage drop across the selected MOSFET BS170, with a maximum drain-source resistance of 5 ohms and a drain or relay current of 83 mA at 0.42 V, is twice as high as the saturated collector-emitter voltage of the T3 in Figure 14. The typical drain-source resistance is only 1.8 ohms. The maximum possible voltage loss of 0.42 V is permissible, unless the relay used really has very tight tolerances in the operating voltage. You should definitely test something like this. If the functioning of the circuit is uncertain, the BS170 can be exchanged for a BSS295 (see sub-figure 9.2).
11. Link list
Thomas Schaerer, August 17, 2004; 09/17/2004; 03/02/2006; October 13, 2006; 02/13/2008; December 17, 2011; 05.12.2013; 08/10/2020;
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